Method and circuit assembly for the resonance damping of stepper motors

ABSTRACT

A method and a circuit arrangement for damping stepper motor resonances during operation of a stepper motor (M), in particular in the medium und high speed range, is described, wherein the coils (A; B) of the stepper motor (M) are each connected into a bridge circuit (Br  1 ; Br 2 ) comprising semiconductor switches (Sw 1 , . . . Sw 4 ), in order to impress into the coils (A; B) a predetermined target coil current (I SollA ; I SollB ). The resonance damping is essentially achieved by activating a passive FD-phase in the zero crossing of the target coil current (I SollA ; I SollB ), during which all 10 semiconductor switches (Sw 1 , . . . Sw 4 ) are opened or switched blocking, in order to thereby feed a coil current flowing in the related motor coil (A; B) back into the supply voltage source either via inverse or body diodes and/or via diodes (D 1 , . . . D 4 ) connected in parallel to the semiconductor switches (Sw 1 , . . . Sw 4 ) in the reverse direction between the positive supply voltage (+V M ) and ground potential.

CROSS REFERENCE TO RELATED PATENT APPLICATIONS

The present application claims priority benefit, to the U.S.Nonprovisional patent application Ser. No. 16/632,360, filed on Jan. 19,2020, entitled “METHOD AND CIRCUIT ASSEMBLY FOR THE RESONANCE DAMPING OFSTEPPER MOTORS”, and listing as inventor Bernhard DWERSTEG, whichapplication is a 371 of PCT/DE2018/100718, filed on Aug. 16, 2018,entitled “METHOD AND CIRCUIT ASSEMBLY FOR THE RESONANCE DAMPING OFSTEPPER MOTORS”, and listing as inventor Bernhard DWERSTEG, whichapplication claims priority benefits to the German Patent ApplicationNo. 10 2017 118 837.1, filed on Aug. 17, 2017, entitled “METHOD ANDCIRCUIT ASSEMBLY FOR THE RESONANCE DAMPING OF STEPPER MOTORS”, andlisting as inventor Bernhard DWERSTEG. Each reference mentioned in thispatent document is incorporated by reference herein in its entirety.

BACKGROUND AND SUMMARY

The invention relates to a method and a circuit arrangement for dampingstepper motor resonances during the operation of a stepper motor,especially in a medium and high speed range.

It is well known that in stepper motors a magnetic rotor is rotated stepby step by a corresponding angle by means of a controlledelectromagnetic field generated by static motor coils and rotating stepby step.

Often there is the desire to be able to rotate the motor with thesmallest possible step angles in order to achieve the highest possibleresolution or accuracy of positioning and a more uniform course of thetorque curve. For this reason, the so-called micro-step mode ispreferred to the known full-step and half-step mode, in which thecurrents flowing through the motor coils are not only switched on andoff, but rise and fall in a certain way. The resolution and uniformitywith which the stepper motor executes the micro-steps depends on howmany different current amplitudes are used to control the motor coilsand how precisely these can be maintained. A sinusoidal or co sinusoidalexcitation of the motor coils is generally the most appropriate, sinceby this microstep-optimized motors can also achieve very continuous,i.e. jerk-free rotation and thus smooth motor running as well as highposition resolution.

For the electrical control of stepper motors, in particular inmicro-step operation, known chopper methods are used, for example, bymeans of which the current direction and current level or current formgiven at any given time by a specified current (target coil current) isimpressed into each of the motor coils by means of PWM current pulseswith the aid of a motor supply voltage (DC voltage) in order to drivethe rotor of the motor with the rotating magnetic field induced thereby.

However, it has been shown that stepper motors can be subject todifferent resonances during operation, which vary greatly depending onthe motor speed and which often cannot be sufficiently suppressed evenwhen the motor is controlled by means of a chopper method.

In particular, undesired motor resonances can occur in the range ofmedium and high speeds and during speed jumps at which the torque of themotor decreases, which resonances can only be suppressed insignificantlyeven in micro-step operation. Furthermore, such resonances, especiallyat constant speed, can also be caused by beatings between speed andchopper frequency or by unfavourable feedback of the counter-EMF inducedin the motor coils to the chopper control.

Even if these resonances generally have only low energy, in theunfavourable case they can build up over many 10 current periods andthen lead to a step loss of the motor in the absence of damping.

These resonances could of course be avoided or reduced by doing withoutthe relevant critical speed ranges or driving through them quickly or bytaking other measures when designing the speed curve. Rubber dampers orsuitable couplings could also be used on the mechanical side. How-ever,all this is considered disadvantageous.

The prevention of resonances by means of encoder-based regulatingprocesses is generally out of the question because the motors concernedare usually operated as purely controlled systems for cost reasons.

For the drive-side reduction of resonances, methods forcounter-EMF-based detection and active damping of resonances or passivedamping methods could be used, in which excess energy from theoscillations is either converted into thermal energy (internal motorresistance) or fed back into the supply voltage source of the motor. Atmedium and high speeds, however, this is not possible or only possibleinsufficiently compared to low speeds, since the electrical commutationfrequency of a stepper motor then reaches a comparable order ofmagnitude of the typical chopper frequency, which is usually slightlyabove 16 kHz.

From the JP H07-95799A it is known that the currents flowing through amotor coil are switched by means of a bridge circuit consisting of foursemiconductor switches and that all semiconductor switches are blockedat a given falling coil current and that the coil current isacceleratedly reduced by diodes connected in parallel to thesemiconductor switches in order to enable a higher motor speed.

Furthermore, the US 2016/0352272 A1 provides a motor controller in whichH-bridge circuits for the motor coils are switched to reduce the powerdissipation in such a way that a coil current to be reduced is fed backinto a power supply.

However, the above-mentioned problem of the formation of motorresonances is not dealt with in these publications.

It is desirable to provide a method and a circuit arrangement with whicha stepper motor can be operated in a relatively simple manner, at leastlargely without resonance, especially in the medium and high speed rangeand especially in micro-step operating mode.

An aspect of the invention involves activating passive fast-decay (FD)phases in the zero crossing of a predetermined target coil current (andpreferably in a region temporally surrounding this zero crossing), withwhich the actual coil current is very quickly reduced by feeding it backinto the supply voltage source and is thereby substantially brought tothe value zero, in order to largely avoid such resonances, in particularin the region of medium and high speeds at which the motor torquedecreases and the motor is therefore particularly sensitive toresonances. This also dampens or suppresses load angle oscillations ofthe motor caused by resonances or speed jumps which lead to acounter-EMF induced in the coils.

Tests have shown that the passive FD phases can also be activated ateach zero crossing of the target coil current independent of a momentarymotor speed, without affecting the actual coil current in an undesirableway in the low or medium speed range.

To activate the passive FD phases, the motor coils are preferably eachconnected in a semiconductor bridge circuit, via which a supply voltage+VM of a supply voltage source is applied to the motor coils, in orderto impress the predetermined target coil current values withpredetermined polarity into the motor coils for each point in time bycorrespondingly opening and closing the semiconductor switches (e.g.MOSFETs). The passive FD phases are then preferably activated by openingor blocking all semiconductor switches in the zero crossing of thetarget coil current (and preferably in a region temporally surroundingthis zero crossing) and thereby feeding back the actual coil currentswhich then still flow either via the inverse or body diodes which areintrinsically existing in the semiconductor switches, and/or via(external) diodes connected in parallel to the semiconductor switches inthe reverse direction between the positive supply voltage +VM andground, into the supply voltage source very quickly, and much fasterthan would be possible with an active FD phase described below.

These passive FD phases can preferably be implemented in combinationwith a chopper method mentioned above, in which the target coil currentvalues preset for each point in time are impressed into the coils byactivating ON-, active FD- and, if necessary, SD-phases. This will beexplained in detail with reference to FIG. 4 .

One advantage of these solutions is that the entire speed range madeavailable by the motor can now be used at least largely resonance-free.

As already mentioned, the inventive solution is preferably used formicro-step operation, but can also be used for any other motor operationin which the coil current is not only switched on and off as infull-step operation, but is also impressed into the coils withincreasing and decreasing current amplitudes or a plurality of currentamplitudes between the value zero and the maximum permissible currentvalue, as for example in half-step, quarter-step, eighth-step operation,etc.

The dependent claims disclose advantageous embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

Further details, characteristics and advantages of the invention resultfrom the following description of preferred embodiments on the basis ofthe drawing. It shows:

FIG. 1 resonances of a stepper motor in full-step operation;

FIG. 2 a speed course which is subject to resonances;

FIG. 3 a mechanical equivalent circuit diagram for a resonant state;

FIG. 4 circuit diagrams of different current flow phases in a motor coilduring chopper operation;

FIG. 5 a schematic diagram of an exemplary circuit arrangement forcarrying out a method according to the invention;

FIG. 6 measured courses of coil current and coil voltages at low speed;

FIG. 7 measured courses of coil current and coil voltages at mediumspeed without motor load;

FIG. 8 measured courses of coil current and coil voltages at mediumspeed with motor load;

FIG. 9 measured courses of coil current and coil voltages at high speedwith resonances; and

FIG. 10 measured courses of coil current and coil voltages at high speedwithout resonances.

DETAILED DESCRIPTION OF EMBODIMENTS

To illustrate the problem underlying the invention, FIG. 1 shows theresonances of a stepper motor in full-step operation. The number of fullsteps S is plotted on the vertical axis and the time t is plotted on thehorizontal axis. The Figure shows that after the execution of a step,the rotor performs resonant oscillations, the amplitudes of which caninitially reach almost to the next or previous step position and thendecay. Such behaviour can also occur in half-quarter-, eighth-stepoperation etc. as well as in micro-step operation and results in asignificant loss of torque and possibly also step losses.

FIG. 2 shows an example of a speed course of a stepper motor, where thespeed v is plotted on the vertical axis and the time t is plotted on thehorizontal axis. From this illustration it becomes clear that even withspeed jumps such as the transition from an acceleration to a constantspeed or number of revolutions of the motor shown here, the oscillationsor resonances mentioned can occur and only gradually subside. Here, too,noticeable torque losses and possibly also step losses can occur.

FIG. 3 shows a mechanical equivalent circuit diagram of this resonantmotor behaviour. The load L of the motor corresponds to a mass m onwhich the torque D of the motor, corresponding to a spring with a springconstant c, acts on the one hand, and a friction R, corresponding to adamping element with a damping constant r, acts on the other hand, sothat a force F(t) is exerted on the mass m which leads to a displacementX(t) of the mass m (corresponding to a motor rotation) in the form of adecaying oscillation.

FIGS. 4(A) to (D) show circuit diagrams with a motor coil A and coilcurrents I_(A) in four different current flow phases.

These Figures show a bridge circuit Br1 from a first to a fourth switchSw1, Sw2, Sw3, Sw4, where a first and a second switch Sw1, Sw2 as wellas a third and a fourth switch Sw3, Sw4 are each connected in series andthe two series connections are connected in parallel. The related motorcoil A is connected to the centres of the bridge branches. The base ofthe bridge circuit is connected to ground via a resistor R_(S1) forsensing the actual motor current flowing through the motor coil, whilethe head of the bridge circuit is connected to the motor supply voltagesource +V_(M). The switches Sw1, Sw2, Sw3, Sw4 are switched by means ofa driver circuit to which chopper switching signals are fed foractivating the current flow phases.

This means that the current actually flowing through the related motorcoil A is measured, and this current is regulated in dependence on themeasurement in positive and negative direction or polarity by means ofcurrent flow or chopper phases (ON, SD, FD) of a chopper method, whichare suitably activated and timed by PWM control of the switches, in sucha way that the actual coil current in each chopper phase and thus overits entire course at least largely corresponds to the course and thepolarity of the related target coil current Isoll.

It is assumed that in the case of a sinusoidal coil current, thepolarity of the coil current is positive in the first and secondquadrants and negative in the third and fourth quadrants.

FIG. 4(A) shows the switch positions and the resulting direction ofcurrent flow I_(A) from the supply voltage source +V_(M) through thecoil A to ground during a first chopper phase (ON-phase) in which thisdirection is the same as the momentary predetermined direction orpolarity of the target coil current, the first and fourth switches Sw1,Sw4 being closed and the second and third switches Sw2, Sw3 being open.

During this ON-phase (positive switch-on phase), the coil current isactively impressed into a coil in the direction of the momentarilyspecified polarity or direction of the coil current, so that the coilcurrent amount increases relatively quickly and continuously (switch-onperiod) until it has reached its momentary target value and the ON-phaseis then terminated. The direction of the coil current impressed by anON-phase is therefore equal to the momentary polarity or direction ofthe coil current.

FIG. 4(B) shows the switch positions and the resulting reversal of thepolarity of coil A as well as the feedback of the coil current I_(A)into the supply voltage source +VM, i.e. contrary to the momentarilygiven direction or polarity of the target coil current (which here isthe same as in FIG. 4(A)) during a second chopper phase (activeFD-phase), in which the first and fourth switches Sw1, Sw4 are open andthe second and third switches Sw2, Sw3 are closed.

During this active FD-phase (negative switch-on phase), the coil currentis actively reduced contrary to the momentarily given polarity of thetarget coil current by reversing the polarity of the coil and feedingthe coil current back into the supply voltage source +VM until it hasreached its momentary target value and the active FD-phase is thenterminated. Alternatively, however, an active FD-phase can also beterminated unregulated after a predetermined period of time has elapsedwith the aim of ensuring the maximum reduction of the coil current in acertain application in the respective active FD-phase on the basis ofempirical values without actually measuring it. In any case, the activeFD-phase serves to reduce the coil current relatively quickly,especially in the phases of decreasing coil current amount (i.e. duringthe second and fourth quadrants of a sinusoidal coil current).

FIG. 4(C) shows a third chopper phase (SD-phase) in which the coil A isshort-circuited or bridged, i.e. the second and fourth switches Sw2, Sw4are closed, while the first and third switches Sw1, Sw3 are open (orvice versa accordingly), so that the current I_(A) flowing in the coil Agradually decays, i.e. recirculates, according to the internalresistance of the coil A.

During this SD-phase (recirculation phase), the related coil is notactively driven but short-circuited or bridged, so that the coil currentdecreases only gradually (i.e. slower than during the FD-phase) due tothe internal resistance of the coil and the counter-EMF. In this phase,the coil current I_(A) cannot usually be measured, so that the SD-phasemust be terminated after a predetermined time period has elapsed,wherein the same constant time period is usually set for all SD-phases.

In summary, these three chopper phases are activated, combined anddimensioned temporally by chopper switching signals generated by thechopper and fed to a driver circuit for the motor coils in such a waythat the actual coil current list is as close as possible to and exactlyfollows the related specified current (target coil current I_(soll)) forthe corresponding motor coil over its entire (e.g. sinusoidal) course,i.e. during the rising and falling coil current phases.

This makes it possible to damp motor resonances sufficiently in mostcases, especially in the low speed range where the counter-EMF of themotor (i.e. the voltage counter-induced in the coils by the rotor) issmaller than the supply voltage.

At medium and high speeds, at which the counter-EMF of the motor reachesor exceeds the level of the supply voltage, the target coil current canoften only be imperfectly impressed into the coils with such a choppermethod, so that the counter-EMF induced in the coils and the associatedoscillations of the motor load angle or motor resonances cannot bedamped by the chopper operation, or cannot be damped sufficiently.However, this can be achieved by activating passive FD-phases as shownin FIG. 4(D).

FIG. 4(D) shows the bridge circuit Br1 comprising the first to fourthswitch Sw1, Sw2, Sw3, Sw4 as well as a first to fourth diode D1, D2, D3,D4, which are connected between the positive supply voltage source +VMand ground in the reverse direction, wherein one diode is parallel toone switch each, and in which all switches Sw1, Sw2, Sw3, Sw4 are open.

If semiconductor switches, in particular MOSFETs, are used as switches,the diodes D1 to D4 are implemented in the form of the inverse or bodydiodes intrinsically existing in the semiconductor switches.Alter-natively or additionally, as shown, external diodes D1 to D4connected in parallel to the switches can also be used.

With this circuit, by opening all switches Sw1, Sw2, Sw3, Sw4 in thezero crossing of the target coil current I_(soll) (and preferably in atemporal range surrounding this zero crossing), a passive FD-phase isactivated in which the coil current I_(A) is reduced via the diodes D4,D1 or D2, D3 and fed back into the supply voltage source +VM.

The duration of the passive FD-phase in the temporal range surroundingthe zero crossing of the target coil current is preferablyexperimentally determined and fixed for a specific motor to becontrolled. It can also be set by a user and, if necessary, adapted to amomentary motor load during motor operation, or remain constantregardless of the motor speed.

The passive FD-phases are preferably activated independently of amomentary motor speed at each zero crossing of the target coil current,i.e. even at low speeds. However, it would also be possible to activatepassive FD-phases only if the motor reaches or exceeds a predeterminedspeed, in particular a medium speed, and/or performs a speed change pertime unit which exceeds a predetermined limit value.

This passive FD-phase has the advantage that in case of a momentarytarget coil current value of zero, the actual coil current I_(A) isreduced particularly quickly to this value of zero. As will be explainedlater in connection with the circuit arrangement, for these passiveFD-phases the semiconductor switches are preferably not opened bydriving the driver circuit Tr by means of the chopper as in the first tothird chopper phase described above, but by a switching signal feddirectly to the driver circuit Tr, so that no delays occur due to thecorresponding generation of the chopper switching signals as in thefirst to third chopper phase.

In addition to the use of the intrinsically existing inverse or bodydiodes and/or the (external) diodes connected in parallel to thesemiconductor switches as described above, this leads to a furtheracceleration of the activation of the passive FD-phases, which isadvantageous for the suppression of motor resonances.

If the motor is subject to load angle oscillations due to resonances orjumps or steps in the drive speed (rotational speed), especially atmedium and high rotational speeds as described above, this also leads toa corresponding variation of the counter-EMF, the level of which canreach and exceed the level of the supply voltage. During the passiveFD-phase, this counter-EMF is also reduced very quickly, so that at thetime of the zero crossing of the target coil current, the zero crossingof the actual coil current is forced or forced very quickly andindependently of the momentary state of the chopper. The reaction timeof the diodes can be neglected since it is in any case considerablyshorter than the reaction time of the first to third chopper phasedescribed above.

Thus, in total, each period of the actual coil current is composed of aplurality of chopper phases as shown in FIGS. 4(A) to 4(C) and thepassive FD-phases as shown in FIG. 4(D), the latter being activated inthe zero crossing region of the target coil current, so that aparticularly precise zero crossing of the actual coil current isachieved and any motor resonances and load angle oscillations areeffectively suppressed.

At low motor speeds, any resonances of the motor are well damped by thechopper phases mentioned, and the passive FD-phases have an almostnegligibly small temporal portion, so that they have no influence on theimpressing of clean sinusoidal current waves by the chopper phases anddo not interfere with motor operation, especially since the passiveFD-phases are only activated when the target coil current is zero oralmost zero. At medium and high motor speeds, the relative temporalportion of the passive FD-phases automatically increases, since the zerocrossings of the coil currents follow each other correspondingly faster,so that any oscillations or resonances of the motor are effectivelydamped.

Since the energy of the oscillations to be damped is relatively lowcompared to the power consumption of the coils, a temporal portion ofthe passive FD-phases of between about 5% and a few 10% of the durationof an electrical period of the target coil current has proven to besufficient to effectively suppress such oscillations, especially at adesired, nominal, or for an application intended, or at a medium ormaximum motor speed.

In the case of a high speed and a high motor load, it may occur that dueto the associated large load angle, the zero crossing of the actual coilcurrent considerably shifts in relation to the zero crossing of thetarget coil current, especially if the motor current cannot becontrolled or reduced quickly enough for example by activating theactive FD-phases (FIG. 4(B)). In this case, the passive FD-phase (FIG.4(D)) activated in the zero crossing of the target coil current has noor only a slight immediate effect. However, if the motor is subject tooscillations, these oscillations are effectively damped by the passiveFD-phases at the moment when the zero crossing of the actual coilcurrent overlaps due to the displacement of the load angle caused by theoscillations, with the zero crossing of the target coil current.

FIG. 5 shows a block diagram of an exemplary embodiment of a circuitarrangement for carrying out the method according to the invention forone of the two coils A, B of a stepper motor.

The circuit arrangement comprises as components known per se anintegrated motor-driver circuit Tr, with which a first bridge circuitBr1 connected between a supply voltage source +V_(M) and ground isdriven via first outputs HS (High Side), LS (Low Side) and BM (BridgeCentre), in order to switch the chopper phases as described above withreference to FIG. 4 , and to impress the currents with correspondingpolarities into a first coil A of the stepper motor M.

The actual coil current I_(A) flowing through the first coil A ismeasured by the voltage drop at a first measuring resistor R_(SA) at thebase of the first bridge circuit Br1.

The second coil B of the motor M (in this example a 2-phase motor) isconnected to a second bridge circuit Br2 with a second measuringresistor R_(SB), which is controlled as described above via secondoutputs HS (High Side), LS (Low Side) and BM (bridge Centre) of thedriver circuit Tr, which are not shown here.

The method according to the invention can also be applied in acorresponding way to a 3-phase or a multi-phase motor with acorrespondingly higher number of motor coils into which the coilcurrents are impressed as explained above using the method according tothe invention.

The components of the circuit arrangement according to the inventionwith which the driver circuit Tr is controlled via its inputs A1, A2 andwhich are described in the following, are only shown for one of the twocoils (namely a first coil A) of the stepper motor M. These componentsmust therefore be implemented once again for the other motor coil B (andpossibly for each additional motor coil in the case of a multi-phasestepper motor) and have to be connected to the corresponding inputs B1,B2, . . . of the driver circuit Tr.

The positive or negative voltage, which drops according to the momentarypolarity of the actual coil current at the measuring resistor R_(SA), isfed to a first input of a comparator K, the second input of which isconnected to the output of a digital-to-analogue converter DAC, withwhich the supplied target coil current values I_(SollA) for the firstcoil A, preferably generated in the digital domain, are converted intoanalogue voltage values, in order to compare a momentary actual coilcurrent value with the momentary target coil current value I_(SollA). Ifthe target coil current values I_(SollA) are supplied in analogue form,the digital-to-analogue converter DAC is of course not required.

As an alternative and corresponding to the detection shown by means of acomparator K, the actual coil currents could also be detected via an ADC(analogue-to-digital converter) in order to carry out the signalprocessing completely in the digital domain.

The output signal at the output of the comparator K is fed to a firstinput e1 of a chopper circuit CH.

A signal PIN indicating the specified direction (polarity) of themomentary target coil current value I_(SollA) is applied to a secondinput e2 of the chopper circuit CH.

The chopper circuit CH generates the chopper switching signals atoutputs a1, a2 as a function of the output signal of the comparator K,for example in the form of PWM pulses Hs, Ls, which are transmitted viaa switching device S to inputs A1, A2 of the driver circuit Tr and withwhich the driver circuit Tr opens and closes the semiconductor switchesof the first bridge circuit Br1 in such a way that, as has beenexplained above with reference to FIGS. 4(A) to 4(C), the actual coilcurrent course flowing through the first motor coil A corresponds atleast substantially to the sup-plied target coil current courseI_(SollA). The same applies accordingly to the second motor coil B andthe control of the second bridge circuit Br2 by means of chopperswitching signals which are fed to the inputs B1, B2 of the drivercircuit Tr.

The circuit arrangement further comprises a detector D for a currentzero crossing, to the input of which the supplied target coil currentvalues I_(SollA) are applied, a timer T, preferably in the form of amono flop, the input of which is connected to an output of the detectorD, and the switching device S having first inputs which are connected tothe outputs a1, a2 of the chopper circuit CH, and a second input(control input), to which a control output of the timer T is applied.The outputs of the switching device S are connected to the inputs A1, AZof the driver circuit Tr.

Thus, if detector D detects a zero crossing of the target coil currentcourse I_(SollA), it generates a trigger signal at its output, whichstarts the timer T. The timer T then generates at its control output acontrol signal preferably having a predetermined duration, for examplein the form of a logical “1” level, which is fed to the switching deviceS. This control signal causes the switching device S to interrupt thechopper switching signals supplied by the outputs a1, a2 of the choppercircuit CH during the presence of the control signal, and instead toapply a zero crossing signal to the inputs A1, AZ of the driver circuitTr, with which all switches (MOSFETs) of the first bridge circuit Br1are opened or are switched to the high-impedance or blocking state sothat, as shown in FIG. 4(D), the passive FD phase for the first coil Aof motor M is activated nearly without any delay and thus substantiallyfaster than would be possible by driving the driver circuit Tr by meansof the chopper circuit CH. The same accordingly applies to the secondand any further coils of the motor.

As soon as the predetermined duration has expired, the timer T switchesoff the control signal again or generates a logical “0” level at itsoutput. Thus the switching device S switches through the chopperswitching signals at the out-puts a1, a2 of the chopper circuit CH againto the inputs A1, AZ of the driver circuit Tr, so that the first bridgecircuit Br1 is controlled again according to FIGS. 4(A) to 4(C).

If these passive FD-phases are each to be activated in a temporal rangesurrounding the zero crossing of the target coil current and thus to beactivated also before the zero crossing is reached, the momentary targetcoil current values I_(SollA) are preferably compared in the zerocrossing detector D with a corresponding threshold value, wherein thetrigger signal is generated when the values fall below this thresholdvalue.

Both this threshold value and the time duration for which the controlsignal is generated at the output of the timer T and by this a passiveFD-phase is activated, can either be given constantly (e.g. for aspecific motor type) or adjustable by a user. In particular, they can beadapted to the motor parameters and/or an actual motor load and/or anactual motor speed.

FIGS. 6 to 10 show temporal oscillograms of the actual (measured) coilcurrent IA and the actual (measured) coil voltages U₁, U₂ at the twoterminals of coil A, each in relation to ground potential, for differentspeed ranges, wherein within the time ranges marked by A1, B1, C1 and D1each a passive FD-phase according to FIG. 4(D) is activated and thus thecoil voltages U₁, U₂ correspond to the respective induction voltage ofthe motor, which is generated in the inductance of the coil or due tothe counter-EMF of the motor during these time periods.

The passive FD-phases can also be recognized by the fact that the coilvoltages U₁, U₂ are not firmly coupled to ground potential or thepositive supply voltage, but increase (first coil voltage U₁) ordecrease (second coil voltage U₂) sharply by the diode voltage with theactivation of the passive FD-phase, and can take intermediate levelsduring the duration of a current-free coil.

Outside the time ranges A1, B1, C1 and D1, the coil voltages U₁, U₂ atthe terminals of the coil result from the switching of the switches Sw1,. . . Sw4 of the bridge circuit Br1 by means of the chopper switchingsignals Hs, Ls generated by the chopper circuit CH according to FIGS.4(A) to 4(C). These time ranges will not be discussed further below.With the oscillograms only the effect of the passive FD-phases indifferent speed ranges and load states of the motor is to be clarified.

FIG. 6 shows the oscillograms at a low motor speed and without or only asmall motor load.

The time range marked with the letter “A” in the upper part of FIG. 6 isexpanded in the lower part. The term “A1” denotes the passive FD-phasewhich is activated by the timer T and which in this case corresponds tothe zero crossing of the actual coil current IA, the zero crossing beingalong the line marked by I_(A)=0.

At the beginning of the time range A1, the actual coil current I_(A) isfed back to the supply voltage source +V_(M). This can be recognized bythe fact that, as mentioned above, the coil voltages U₁ and U₂ are eachslightly above or below the voltage range (namely that of the supplyvoltage) visible in previous chopper cycles. The difference ofapproximately 0.7 Volt is due to the diode flux voltage of the MOSFETswitches Sw1, . . . Sw4 used in the bridge circuit Br1. The feedbackdecays within the time range A1 of the passive FD-phase, so that theactual coil current I_(A) becomes zero and the coil voltages U₁, U₂show, after initial feedback, only a small difference due to thecounter-EMF. The termination of the passive FD-phase is shown in theoscillograms by the steep drop of the two coil voltages U₁ and U₂.

FIG. 7 shows the oscillograms of the measured coil current IA and of thecoil voltages U₁, U₂ measured at the terminals of coil A at a mediummotor speed and without a motor load. The time range marked with theletter “B” in the upper part of FIG. 7 is shown expanded in the lowerpart.

The term “B1” refers to the passive FD-phase activated by the timer T,where the start of an FD-phase leads to the jump points in the course ofthe coil voltages U₁, U₂ marked with a dotted line d, and the end of thepassive FD-phase results in a steep drop in the coil voltage U₁.

In this case, the passive FD-phases have no influence on the actual coilcurrent I_(A), the zero crossing of which, due to the very small loadangle in the measured situation, is only slightly lagging behind thetarget coil current I_(SollA) and is only reached after the end of thepassive FD-phase (time range B). This illustration makes it particularlyclear that the passive FD-phase not only has no influence on the coilcurrent I_(A), but also no interference occurs.

FIG. 8 shows the oscillograms of the measured coil current IA and of thecoil voltages U₁, U₂ measured at the terminals of coil A, also at amedium motor speed, but with motor load and a resulting displacement ormagnification of the load angle, wherein the start of the FD-phase againleads to the jump points in the course of the coil voltages U₁, U₂marked with a dotted line d, and the end of the passive FD-phase resultsin a steep drop in the coil voltage U₁.

The time range marked with the letter “C” in the upper part of FIG. 8 isagain shown expanded in the lower part.

Within the time range C1, i.e. within the passive FD-phase activated bythe timer T, the actual coil current I_(A) is partially fed back to thesupply voltage source, and thus an actually activated attenuation takesplace.

In addition, by activating the passive FD-phases in the zero crossing ofthe target coil current I_(SollA), the actual coil current I_(A) is muchmore pronounced and reaches the zero current in a more defined way.

FIG. 9 shows the oscillograms of the measured coil current I_(A) and thecoil voltages U₁, U₂ measured at the terminals of coil A at a high motorspeed without activating passive FD-phases.

In this case, the motor is in a resonant oscillation state, wherein thecourse of the coil current I_(A) is distorted due to the associated loadangle oscillations and differs from cycle to cycle due to theoscillations. In the worst case, this operating state can lead to aresonance catastrophe.

In comparison to this and in the same scales, FIG. 10 shows theoscillograms of the measured coil current I_(A) and the coil voltagesU₁, U₂ measured at coil A again at a high motor speed, but withactivated passive FD-phases.

The term “DI” indicates the range of a passive FD-phase activated by thetimer T. By this and in comparison to the operating state shown in FIG.9 , a large part of the oscillation energy is absorbed, and the coilcurrent I_(A) only has considerably lower amplitudes and reaches a zerocurrent much more pronounced and defined.

What is claimed is:
 1. A method for damping stepper motor resonances,the method comprising: coupling semiconductor switches of a bridgecircuit with a plurality of motor coils; applying a voltage from avoltage source to the plurality of motor coils, the voltage havingpredefined current values and a predefined polarity, by correspondinglyopening and closing the semiconductor switches; and activating a passivefast-decay phase (FD-phase) at stepper motor speeds in a zero crossingor in a time range surrounding the zero crossing of a predefined targetcoil current for each motor coil within the plurality of motor coils,the passive FD-phase causes a coil current to flow from a first motorcoil within the plurality of motor coils back into the voltage source bydiodes electrically coupled with the semiconductor switches in thereverse direction between the positive supply voltage and groundpotential.
 2. The method according to claim 1 wherein the passiveFD-phase is activated in response to a counter of the stepper motorcrossing a voltage level applied by the voltage source.
 3. The methodaccording to claim 1 wherein the passive FD-phase is activated for aperiod of time between about 5% and 10% of the temporal duration of anelectrical period of the target coil current values.
 4. The methodaccording to claim 1, wherein the passive FD-phase is activatedindependently of the stepper motor speed.
 5. The method according toclaim 1 wherein the passive FD-phase is activated when the stepper motorhas a speed change that exceeds a predetermined value.
 6. The methodaccording to claim 1, wherein the diodes comprise body diodesintrinsically existing in the semiconductor switches and/or diodescoupled electrically in parallel with the semiconductor switches.
 7. Themethod according to claim 1 wherein the passive FD-phase is activated bya switching signal which is generated in response to the zero crossingor a temporal range surrounding the zero crossing of the target coilcurrent values.
 8. A device for damping stepper motor resonances, thedevice comprising: a bridge circuit comprising a plurality ofsemiconductor switches electrically coupled to a plurality of motorcoils; a supply voltage coupled to the bridge circuit, the supplyvoltage provides a voltage to the motor coils to generate target coilcurrent values by opening and closing the semiconductor switches; adetector coupled to detect a zero crossing of the target coil currentvalues for at least one motor coil within the plurality of motor coils;a timer coupled to the detector, the detector compares the target coilcurrent values with a predetermined threshold value, the timer beingactivated when at least one of the target coil current values fallsbelow a predetermined threshold value, and a switching device coupled tothe plurality of semiconductor switches and the plurality of motorcoils, the switching device controls the plurality of semiconductorswitches to causes a coil current to flow from a first motor coil withinthe plurality of motor coils back into the voltage source by thesemiconductor switches, the coil current flowing from the first motorcoil into the voltage source for a period of time determined by thetimer.
 9. The device of claim 8 wherein the plurality of semiconductorswitches comprises MOSFETs.
 10. The device of claim 8 wherein the periodof time determined by the timer is predetermined or user-adjustable. 11.The device of claim 8 wherein the timer comprises a mono-flop.
 12. Thedevice of claim 8 wherein the plurality of semiconductor switchescomprises a plurality of diodes, the plurality of diodes causes the coilcurrent to flow from a first motor coil within the plurality of motorcoils back into the voltage source.
 13. The device of claim 12 whereinthe plurality of diodes comprises body diodes intrinsically existing inthe semiconductor switches, the plurality of diodes.
 14. The device ofclaim 8 wherein a plurality of diodes is coupled electrically inparallel with the plurality of semiconductor switches, the plurality ofdiodes causes the coil current to flow from a first motor coil withinthe plurality of motor coils back into the voltage source.